Antennas collect radio waves and channel them into radio frequency (RF) transmission lines which direct the signals to circuits from which information can be demodulated and decoded. Glass, the most common portal between outside and inside environments, is clear at the visible part of the electromagnetic spectrum, and it is also relatively transparent to a large portion of the electromagnetic spectrum useful for radio wave communications. Since glass as a building material is used everywhere, it could be a readily accessible substrate upon which to mount or fabricate the antennas and RF electronics, but only if these circuit components are also transparent. In this paper, we present our development to date of glass RF circuits along two tracks: 1) transparent antennas and 2) graphene based active and passive circuit elements. Along the first track we have demonstrated antennas made from nanowire films capable of an optical transparency of 72% and sheet resistance of 4-5Ω/sq. Along the second track, we have in so far demonstrated graphene on glass field effect transistors with an fmax of 7 GHz, varactors with 1.4:1 tuning range, resistors with 3-70 kΩ, and capacitors from 13-860 pF. This is just the start; our plans are to increase the frequency and tuning ranges of the active and passive devices. Since graphene is inherently transparent at visible wavelengths, we ultimately would like to merge these two tracks to integrate active and passive RF circuitry with the antenna either directly on glass or as an applique put on glass, circuits which we’ve termed RF Glass®.
A W-band unamplified direct detection radiometer module is described that provides a wideband response and is
scalable to large arrays. The radiometer design is intended to provide sufficient sensitivity for millimeter wave imaging
applications with a goal of 2K noise equivalent temperature difference (NETD) at a 30 Hz frame rate.
This effort leverages previously reported device scaling to increase sensitivity. We present a radiometer module
designed for 60 GHz RF bandwidth that utilizes HRL's antimonide-based backward tunnel diode. An impedance
matching circuit with on- and off-chip elements, as well as ridged waveguide, provides a wideband match to the
detectors. Modules were designed with two different microwave substrates: 125 micron thick quartz and 100 micron
thick alumina. flip-chip bonding of the detectors is amenable to automated pick-and-place for high volume
manufacturing. The modular nature of the array approach allows large arrays to be manufactured in a straightforward
manner. We present the design approach along with both electromagnetic simulations and measured performance of the
modules. This work was supported by phase II of DARPA's MIATA program.
We are developing a passive W-band millimeter wave imaging array that operates without the use of RF low noise
amplifiers. The work is supported by the DARPA MIATA program. Previously reported Phase I results were a noise
equivalent temperature difference (NETD) of 4.8°K. The goal of Phase II, currently underway, is to decrease this to 2°K
or less. There are two improvements that must be made to achieve the goal. The square law diode detector sensitivity
and the RF bandwidth reaching the detector must be increased significantly. This paper mainly deals with the first issue,
the effort to increase the sensitivity by decreasing the diode area and capacitance, using electron beam lithography.
Brief mention will be made of the redesign of the antenna-to-diode transition that simulations indicate will provide a
doubling of bandwidth from 30 to 60 GHz.
We have demonstrated a high efficiency package for zero bias Sb-based backward tunnel diodes developed for passive millimeter wave imaging. Flip-chip mounting of detector MMICs onto quartz substrates permit placement of the detector directly within the WR-10 waveguide feeds for diagonal horn antennas. This arrangement minimizes the losses between the detectors and antennas while providing an impedance match over a majority of W band. A 2x2 array of radiometers was fabricated, assembled, and measured using coherent measurement techniques. The resulting noise equivalent temperature difference, calculated assuming a 30 Hz frame rate is 10 degrees K.
We have fabricated a > 10 GHz high speed optoelectronic (OE), selector switch with high isolation (> 57 dB). This switch was used as a receiver for a microwave network demonstration with simulated satellite feeds of FM video and microwave BPSK digital channels.
A universally accepted figure of merit for analog microwave transmission links is the spur-free dynamic range (SFDR), which is the ratio (usually expressed in dB) of the largest to smallest signal a link can transmit and receive without introducing any measurable distortion. This paper presents the result of a study of broadband, microwave fiber optic links that contain high- linearity integrated-optic modulators. This study focused on two distinct modulator forms, the dual-parallel Mach-Zehnder modulator and the linearized directional-coupler modulator. Computer simulations were performed to determine how the SFDR was affected by variations of modulator parameters. In addition, the dynamic range and noise figure of links that included preamplifiers were calculated. The results of the link analysis was that the linearized directional-coupler modulator provided the highest SFDR for a broadband microwave fiber optic link.
A microwave fiber optic link is developed for operation from 8 to 18 GHz. Modulation of a 1.3-μm wavelength optical carrier is accomplished using a lithium niobate Mach-Zehnder interferometer modulator that was designed for broadband operation up to 19 GHz. The link transmitter components, which consist of a high-power semiconductor laser, an optical isolator, and a modulator with polarization-preserving fiber pigtails, are spliced together to fit into a 10.16- x 5.08- x 2.54-cm3 (4- x 2- x 1-in.) box for applications where space is critical. The average link electrical gain is 45 dB, the frequency response is flat to within ±4 dB across the 10-GHz band, and the spur-free dynamic range of the link is 84 dB.
The frequency response of integrated optic modulators with traveling wave electrodes is limited by the phase velocity mismatch between the optical and RF signals. At millimeter wave frequencies the short electrodes that must be used to avoid this phase mismatch severely reduce the modulator sensitivity. One approach that provides high frequency modulators with increased sensitivity is the use of periodic intermittent interaction (PII) electrodes and enhanced periodic phase reversal (EPPR) electrodes. In this paper, the RF signal on the electrode is represented as a summation of traveling wave space harmonics. The electrode geometry is adjusted so that the phase velocity of the dominant space harmonic equals the phase velocity of the optical signal at the design frequency. General velocity-matching conditions are derived for both the PII and the EPPR modulators; the bandpass frequency response is then calculated by adding the individual responses of all space harmonics traveling down the electrode. Design examples are provided of periodic electrodes for LiNbO3 and GaAs modulators for operation at 35 GHz. Finally, a practical method for velocity-matching is described that requires only the two-port characterization of an individual electrode unit section.
Partitioning the transmission line on a LiNbO3 traveling wave modulator into short sections and feeding each section from an antenna integrated to the modulator surface can overcome the frequency limitation from material dispersion. A phase modulator at 5-13 GHz with a peak sensitivity better than 100 deg/(Watt) sq rt is demonstrated using an unoptimized microwave feed system. The technique should be scalable to high millimeter-wave frequencies.
A millimeter wave integrated lithium niobate modulator, consisting of a titanium diffused optical waveguide Mach-Zender interferometer and a traveling wave coplanar waveguide electrode with periodic series stubs, is analyzed through the application of Floquet's theorem. First, a design equation for the modulator is derived by expanding the RF signal along the optical waveguide into space harmonics and then matching the velocity of the dominant space harmonic to the velocity of the optical signal. Then, the frequency response of the modulator is found by integrating, over the modulator's length, the local optical phase shifts that are electro-optically induced by all of the RF space harmonics. Finally, it is shown how the concepts developed here for an ideal (no reflections) periodic structure can be applied to the experimental determination of the modulator response by characterizing isolated unit sections of a real electrode. This approach has the added advantage of facilitating RF inmpedance matching to the modulator.